LPI/LPD communication systems

ABSTRACT

In a wireless communication system, a secure communication link is provided by selecting a decoy data signal for transmission, generating a precoding matrix from a message to be sent; and multiplying the decoy data signal by the precoding matrix to produce a precoded signal. A clean version of the decoy data may be transmitted to an intended receiver. The receiver can distinguish between the information-bearing precoding and the natural random channel distortions of the transmission medium to decrypt the information, while an eavesdropper would find it difficult to distinguish between the natural channel distortions and information-bearing precoding in the signals it receives.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a Continuation of U.S. patent application Ser. No.14/498,499, entitled “Sharing Resources Between Wireless Networks,”filed Sep. 26, 2014, which is a Divisional of U.S. patent applicationSer. No. 13/757,032, entitled “LP/LPD Communication Systems,” filed onFeb. 1, 2013, now U.S. Pat. No. 8,929,550, which claims priority under35 U.S.C. 119(e) to U.S. Provisional Application Ser. No. 61/594,086,entitled “LPI/LPD Communication Systems,” filed Feb. 2, 2012.

BACKGROUND OF THE INVENTION

I. Field of the Invention

The present invention relates generating and distributing a secret keyfor symmetric encryption by employing channel characteristics of thecommunication channel between a pair of transceivers.

II. Description of the Related Art

While a wireless communication link is vulnerable to interception byunintended receivers, the physical characteristics of the propagationchannel (such as reciprocity, highly variable channel distortions, andthe uniqueness of those distortions in any given link) can be exploitedfor generating and exchanging encryption keys, and encryptingtransmissions.

Techniques for key generation commonly employ random processes. Thephysical wireless channel provides the required statistical randomness,and channel reciprocity (i.e., radio waves propagating in bothdirections of a radio link between a pair of transceivers experience thesame multipath channel distortions if their frequencies are the same)potentially enables secure key exchange without transmitting keyinformation that can be intercepted by an unintended receiver.Specifically, each of a pair of transceivers observes a random process(i.e., their propagation channel) that is observed differently by anyunintended receiver. For example, each transceiver estimates the commonchannel from known training signals received from the other transceiver.Temporal and spatial variations of the channel are random. Thus, fortransceivers that are sufficiently distant from each other (e.g.,typically a few wavelengths), their channel transfer functions areuncorrelated.

Practical examples of where reciprocity can be achieved include wirelesssystems employing time-division duplex (TDD), such as 802.11, 802.16(WiMAX), and LTE. However, some aspects of the invention provide forfull-duplex operation in non-TDD systems by cancelling transmit signalleakage into the receiver, which enables a pair of transceivers tosimultaneously transmit and receive signals in the same frequency band.

SUMMARY

The foregoing is a summary and thus contains, by necessity,simplifications, generalizations and omissions of detail; consequently,those skilled in the art will appreciate that the summary isillustrative only and does not purport to be limiting in any way. Otheraspects, inventive features, and advantages of the devices and/orprocesses described herein, as defined solely by the claims, will becomeapparent in the non-limiting detailed description set forth herein.

In one aspect of the invention, at least one node in a first networkrequests a communication channel from a second network. Upon receiving achannel assignment from the second network, nodes in the first networkemploy the assigned channel for communicating in the first network in amanner that is transparent to the second network. The first network maybe a peer-to-peer network and the second network may be a cellularnetwork.

In accordance with one aspect, nodes communicating in the first networkcomply with the second network's data transmission format. For example,the nodes may communicated the first network's network control functionsin predetermined portions of the data payload portion of the secondnetwork's frame format.

In an aspect of the invention, a first node and a second node requestuplink and downlink channels from the second network. The first nodesends a request to the second node to establish a peer-to-peer linkeither before or after receiving assigned channels from the secondnetwork. Upon receiving an acknowledgement from the second node, atleast one of the uplink and downlink channels is employed forcommunicating in the first network. For example, the first node maycommunicated directly with the second node, wherein the “directly” meansthat there are no intervening servers or routers that are part of thesecond network's infrastructure.

In another aspect, a transmitter selects a decoy data signal as acarrier signal, synthesizes channel distortions bearing data; anddistorts the carrier signal with the channel distortions prior totransmission. This method may comprise providing an undistorted versionof the decoy data to an intended receiver. For example, a clean (i.e.,undistorted) version of the decoy data may be transmitted to theintended receiver via an alternative channel in the first network or viathe second network. The decoy data signal may be a retransmittedbroadcast signal or a predetermined data sequence that is stored inmemory or that can otherwise be locally reproduced by the intendedreceiver.

Another aspect of the invention comprises receiving a decoy data signalhaving synthesized data-bearing channel distortions and natural channeldistortions; receiving a clean decoy data signal; and distinguishingbetween the synthesized data-bearing channel distortions and the naturalchannel distortions.

In another aspect of the invention, a transmitting node selects a decoydata signal vector for transmission, generates a MIMO precoding matrixfrom a message to be sent between the transmitting node and a receivingnode; and multiplies the decoy data signal vector with the MIMOprecoding matrix for constructing a precoded signal vector, which istransmitted.

In one aspect, the MIMO precoding matrix comprises transmit power valuesfor each spatial subchannel based on the message and calculatedeigenvalues of matrix HH^(†), where H is the estimated MIMO channelmatrix.

Although particular aspects and embodiments are described herein, manyvariations and permutations of these embodiments fall within the scopeof the invention. Although some benefits and advantages of aspects ofthe invention are mentioned, the scope of the invention is not intendedto be limited to particular benefits, uses, or objectives. Rather,aspects of the invention are intended to be broadly applicable todifferent applications, system configurations, networks, and devices,some of which are illustrated by way of example in the figures and inthe following description. The detailed description and drawings aremerely illustrative of some aspects of the invention rather thanlimiting, the scope of the invention being defined by the appendedclaims and equivalents thereof.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention is illustrated in the figures of the accompanying drawingswhich are meant to be exemplary and not limiting, in which likereferences are intended to refer to like or corresponding parts, andwherein:

FIG. 1 is a flow diagram of a secure communication method that can beconfigured in accordance with aspects of the invention.

FIG. 2 is a block diagram of a wireless communication system that maycomprise components configured in accordance with aspects of theinvention.

FIG. 3 is a flow diagram of a communication method according to anaspect of the invention.

FIG. 4 is a flow diagram for initiating a peer-to-peer communicationlink in accordance with an aspect of the invention.

FIG. 5A illustrates part of a method configured in accordance with anaspect of the invention.

FIG. 5B illustrates part of a method configured in accordance with anaspect of the invention.

FIG. 6A illustrates part of a method configured in accordance with anaspect of the invention.

FIG. 6B illustrates part of a method configured in accordance with anaspect of the invention.

FIG. 6C illustrates part of a method configured in accordance with anaspect of the invention.

FIG. 7A depicts an aspect of the invention wherein two nodes employ thesame third-party downlink assignment for peer-to-peer communications.

FIG. 7B depicts an aspect of the invention wherein a transmitting nodeemploys a third-party broadcast channel for transmitting signals in apeer-to-peer communication link.

FIG. 8 is a flow diagram of a method according to one aspect of theinvention that employs a combination of antenna cancellation, analogcancellation, and digital cancellation for full-duplex operation.

FIG. 9 is a block diagram of a full-duplex transceiver configured inaccordance with an aspect of the invention.

FIG. 10A depicts an aspect of the invention wherein a node employsbeamforming to reduce the SNR of its peer-to-peer transmit signalreceived at a third-party network's receiver.

FIG. 10B depicts an aspect of the invention wherein a first node employsbeamforming to reduce the power of a decoy signal or reference signal ina spatial subchannel used for communicating with a second node.

FIG. 11 is a flow diagram of a method employing adaptation of acooperative-MIMO system according to aspects of the present invention.

FIG. 12A is a block diagram of an OFDM transmitter configured inaccordance with certain aspects of the invention.

FIG. 12B is a block diagram of an OFDM receiver configured in accordancewith certain aspects of the invention.

FIG. 13 is a flow diagram depicting a method for generating a secret keyfrom channel measurements in accordance with aspects of the invention.

FIG. 14A is a block diagram of a transmitter configured in accordancewith aspects of the invention for transmitting known symbols in timeslots and/or subcarrier frequency bins determined from a function of thesecret key.

FIG. 14B is a block diagram of a receiver configured for performingchannel estimation in accordance with an aspect of the invention.

FIG. 15 is a flow diagram depicting a communication method in accordancewith an aspect of the invention.

FIG. 16A depicts an aspect of the invention wherein a third-partytransmission signal is employed as a reference signal.

FIG. 16B depicts an aspect of the invention wherein a first nodetransmits a clean reference signal via a third-party network and adistorted reference signal with information-bearing synthesized channeldistortions via a peer-to-peer channel to a second node.

FIG. 17A depicts an aspect of the invention wherein a node employs afirst peer-to-peer channel for transmitting a clean reference signal anda second peer-to-peer channel for transmitting an information-bearingreference signal.

FIG. 17B depicts an aspect of the invention wherein a destination nodegenerates its own copy of a reference signal.

FIG. 18 depicts a method for providing a communication link with LPI/LPDproperties.

FIG. 19A illustrates a bidirectional channel estimation process in whicheach of a pair of nodes transmits a known training sequence or pilotsignal that is used to estimate their reciprocal peer-to-peer channel.

FIG. 19B illustrates an aspect of the invention wherein a first nodeemploys an additional beam-forming matrix for transmitting an uplinksignal assigned by a third-party network.

FIG. 20 is a block diagram of a transmitter configured in accordancewith an aspect of the invention.

FIG. 21 is a block diagram of a receiver configured in accordance withan aspect of the invention.

FIG. 22A is a flow diagram of a communication method according to oneaspect of the invention.

FIG. 22B is a flow diagram of a communication method according to anaspect of the invention.

DETAILED DESCRIPTION

In the following description, reference is made to the accompanyingdrawings that form a part hereof, and in which is shown by way ofillustration specific aspects in which the invention may be practiced.It is to be understood that other aspects and embodiments may beutilized, and structural changes may be made without departing from thescope of the present invention.

FIG. 1 is a flow diagram of a secure communication method that can beconfigured in accordance with aspects of the invention. A secret key isshared 91 between a first node and a second node. As used herein, asecret key is shared between a pair of legitimate nodes, and istypically kept secret. Key distribution is an important aspect formaintaining security in a cryptographic system, and it is performed in away that minimizes the chance of it being intercepted by an unintendedparty.

The first node employs the secret key in an encryption algorithm forencrypting plaintext data 92. As used herein, plaintext is a message tobe sent between at least two legitimate nodes. Plaintext is an input ofthe encryption process 92. As used herein, the encryption algorithmemploys the secret key for encoding the plaintext to produce acipher-text output. The resulting cipher text is transmitted to thesecond node 93. The second node employs the secret key in a decryptionalgorithm for decoding the cipher text 94, thereby producing an estimateof the plaintext data. As used herein, a decryption algorithm employsthe secret key for decoding the cipher text, which is an encryptedversion of the plaintext.

FIG. 2 is a block diagram of a wireless communication system that may beconfigured to operate in accordance with aspects of the invention. Afirst node (Node 1) 101 and a second node (Node 2) 102 communicate witheach other over a wireless link. Reciprocal channel vectors between Node1 and Node 2 are represented by h_(a) and h_(a′), respectively. Channelsh_(b) and h_(c) convey information to a third, unintended receiver 105(e.g., an eavesdropper).

Due to noise and estimation errors, the channel estimates at Nodes 1 and2 are ĥ_(a)=h_(a)+ε₁ and ĥ_(a′)=h_(a′)+ε₂, respectively, where ε_(i) isa zero-mean complex Gaussian estimation error at node i having varianceσ_(i) ². The channel vectors may represent frequency bins in an OFDMsystem or stacked subspace elements in a MIMO channel. In some aspectsof the invention, a MIMO-OFDM system may be employed.

The size of the secret key depends on the number of unique informationbits that can be extracted from the channel measurements, which dependson the richness of the multipath channel. The theoretical number ofunique bits is expressed byI _(K)=log₂ |R _(aa) R _(σ) ⁻¹ +I|,where R_(σ)=(σ₁ ²+σ₂ ²)I+σ₁ ²σ₂ ²R_(aa) ⁻¹, and R_(aa)=E{h_(aa)h_(aa)^(H)} is the covariance matrix of h_(a).

Since differences in channel estimates at Nodes 1 and 2 can occur due tonoise, the number of unique information bits that can be generated istypically well below the theoretical limit. Therefore, some aspects ofthe invention provide for combining correlated channel samples toprovide each channel measurement with an improved SNR. For example,samples of closely spaced frequency bins, multiple samples within thechannel's coherence time, and/or samples of highly correlated spatialchannels may be combined to produce each channel measurement, thusincreasing the number of unique information bits that can be generatedin practical systems.

If the size of the plaintext exceeds the key size, one aspect of theinvention reduces the amount of data that needs to be encrypted byemploying a data-sanitization technique. Specifically, the amount ofplaintext that requires encryption may be reduced by separatingsensitive data from non-sensitive data in the plaintext information tobe transmitted, followed by encrypting the sensitive data portion. Atiered encryption process may be employed whereby a higher level ofsecurity is provided for encrypting the sensitive data and a less secureencryption may be applied to the non-sensitive data.

FIG. 3 is a flow diagram of a communication method according to anaspect of the invention. At least one of a pair of nodes exploits athird-party communications network, such as a cellular network, forsetting up a peer-to-peer link 301 between the nodes. Once the nodesagree upon radio channels assigned by the third-party network, the nodesmay employ the agreed-upon channels for establishing a full-duplexpeer-to-peer link 302 that is transparent to the third-party network.For example, each of the nodes may simultaneously transmit and receiveradio signals in the same frequency band while employing interferencecancellation for canceling transmit leakage signals in the receiver. Insome aspects of the invention, the nodes may transmit and receivesignals using downlink and/or broadcast channels employed by thethird-party network. In such aspects, full-duplex operation 302comprises employing interference cancellation for cancelling anythird-party network transmissions in the signals received from thepeer-to-peer link.

Each of the nodes may comprise multiple antennas, such as by employingMIMO and/or cooperative-MIMO. In aspects of the invention wherein thenodes employ antenna-array processing (e.g., virtual array processing),any combination of array-processing operations may be adapted 303, suchas the number of antennas, precoding operations, and/or the type ofcombining algorithms employed. In one aspect of the invention, abeaconing protocol may be employed by one or both nodes for determininga preferred set of beam patterns for use in the peer-to-peer link.Adapting the MIMO operation 303 may comprise nulling interference, suchas downlink signals transmitted by the third-party network or jammingsignals transmitted by hostile parties. In some aspects, adapting theMIMO operation 303 comprises beamforming operations that degrade the SNRof unintended receivers (e.g., eavesdroppers).

Channel measurement 304 (e.g., channel estimation) of the peer-to-peerradio channel is performed by both nodes, and a secret key is generatedby each node 305. Various techniques may be employed for improving theprobability of key agreement during either or both steps 304 and 305.

Since the radio channel typically varies with respect to time, channelmeasurements may be updated 306 at predetermined intervals or atintervals determined by the channel measurements or measured bit-errorrates. In some aspects of the invention, channel updates 306 arecoordinated based on the secret key. The timing for the transmission ofknown training symbols or pilot signals may be based on the secret key.In an OFDM system, predetermined symbols (e.g., pilots) may betransmitted on subcarriers denoted by some function of the secret keythat is known by both nodes. In a MIMO system, training symbols may betransmitted in predetermined spatial subchannels relative to somefunction of the secret key.

In various aspects of the invention, the nodes comply with the datatransmission format of the third-party network and employ predeterminedportions of the data payload of a frame for peer-to-peer network controlfunctions. In some aspects, a portion of the radio resource allocatedfor data may be employed by the nodes for channel sounding. Theselection of the subchannels and the timing of the training signals inthe selected subchannels may be determined by the secret key.

In some aspects of the invention, the plaintext data may be modified,such as separated into different levels of sensitivity, which areencrypted using different levels of security. This can effectivelyreduce the data rate to match a lower key rate 307. For example,adapting the data 307 may comprise performing a form of datasanitization of the plaintext to separate sensitive data fromnon-sensitive information. The sensitive data may be encrypted using thesecret key, and the non-sensitive information may be encrypted using analternative mechanism.

A reference signal is selected 308 as a carrier signal for the encrypteddata. The reference signal may comprise a retransmitted third-partybroadcast, a predetermined data sequence that can be locally reproducedby the receiving node, or a decoy signal comprising a “clean” versionsupplied to the receiving node via an alternative channel.

Channel distortions are synthesized 309 based on a function of thesecret key. The channel distortions are imparted to the referencesignal, which is transmitted 310 to the receiving node.

Step 301: Leveraging a Third-Party MAC

FIG. 4 is a flow diagram for initiating a peer-to-peer communicationlink in accordance with an aspect of the invention, such as may beperformed with respect to Step 301 shown in FIG. 3. In this aspect ofthe invention, a pair of nodes employs a third-party network to assign acommunication channel, which the nodes use for peer-to-peercommunications by a means that is transparent to the third-partynetwork. For example, in a cellular network, uplink and/or downlinkmultiple access channels assigned to either or both nodes for cellularcommunications may be employed by the nodes for peer-to-peercommunications. This provides a contention-free radio channel for theduration of the peer-to-peer link and essentially outsources many of theMAC functions required for establishing and maintaining a session.

A first node initiates a communication link with a second node via athird-party network 401. FIG. 5A illustrates a request message x₀transmitted via a third-party network 103 from the first node 101 to thesecond node 102. In another aspect of the invention, the first node 101may transmit the request directly to the second node 102, such as in apeer-to-peer network, and then the pair of nodes 101 and 102 mayestablish a link via the third-party network 103.

In response to the first node's 101 request for a communication link,the third-party network 103 reserves communication resources (e.g.,radio channels) for the link and assigns those resources to the firstnode 101. For example, as illustrated in FIG. 5B, the third-partynetwork 103 assigns a first uplink channel f_(up)(1) and a firstdownlink channel f_(dn)(1) to the first node 101. Similarly, thethird-party network 103 assigns a second uplink channel f_(up)(2) and asecond downlink channel f_(dn)(2) to the second node 102.

Once the nodes are connected via the third-party link, the first nodesends a request to the second node to establish a peer-to-peer link 402.Such requests and other peer-to-peer network control operations may becommunicated inside predetermined portions of the data payload in thethird-party transmission frame format.

The second node 102 responds to the request with an acknowledgement 403(such as depicted in FIG. 6A), which may be returned via the third-partynetwork 103. The acknowledgement may comprise a message x₁ transmittedvia the third-party network. Alternatively, the acknowledgement may betransmitted directly to the first node 101, such as in a peer-to-peernetwork.

The nodes employ the third-party network for exchanging their respectivechannel assignments 404, such as depicted in FIG. 6B. The nodes 101 and102 agree upon at least one of the radio channels assigned by thethird-party network 103. The selected radio channels may comprise anycombination of at least one of the assigned uplink channels, at leastone of the assigned downlink channels, and/or at least one broadcastchannel, such as a broadcast channel employed by the third-partynetwork. Alternatively, the nodes 101 and 102 may exchange their clientinformation (e.g., their client identifications used by the third-partynetwork 103) and then listen for each other's channel assignmentsbroadcast by the third-party network 103.

The nodes 101 and 102 perform a peer-to-peer link set-up 405 using theagreed-upon radio channel(s). For example, FIG. 6B depicts an aspect ofthe invention wherein the first node 101 employs its third-party uplinkchannel for peer-to-peer communications with the second node 102, andthe second node 102 employs its third-party uplink channel tocommunicate with the first node 101 via the peer-to-peer link. In thisaspect, multiple access in the peer-to-peer link is established by thechannel assignments made by the third-party network 103.

Link set-up 405 typically comprises performing signal acquisition. Forexample, the first node 101 transmits in its assigned uplink channel,and the second node 102 determines whether it detects the first node's101 transmission. The nodes 101 and 102 may communicate with each othervia the peer-to-peer link and/or via the third-party network 103 fornetwork control functions, such as beam-forming and/or power control.

FIG. 6C depicts an aspect of the invention wherein both nodes 101 and102 employ the same third-party channel assignment for peer-to-peercommunications. In such aspects, multiple access of the peer-to-peernetwork is controlled by the nodes 101 and 102. For example, the nodes101 and 102 may partition the assigned channel into sub-bands, timeslots, or CDMA channels for multiple access. Alternatively, the nodes101 and 102 may simultaneously transmit in the same band while employingself-interference cancellation to achieve full-duplex operation. Suchsimultaneous transmissions degrade the SNR of signals received by anunintended receiver and make it difficult for the unintended receiver toaccurately estimate its channel.

FIG. 7A depicts an aspect of the invention wherein both nodes 101 and102 employ the same third-party downlink assignment for peer-to-peercommunications. In such aspects, the nodes 101 and 102 must suppressinterference caused by the downlink transmission of the third-partynetwork 103. For example, the nodes 101 and 102 may employ beam-formingto null the third-party downlink transmissions during peer-to-peercommunications.

FIG. 7B depicts an aspect of the invention wherein the first node 101employs a third-party broadcast channel for its peer-to-peercommunication link. In such aspects, the second node 102 may employbeamforming to null the received broadcast from the third-party networkwhile the first node 101 is transmitting.

Step 302: Full Duplex Operation

With respect to Step 302 in FIG. 3, some aspects of the inventionprovide for configuring the nodes 101 and 102 for simultaneouslytransmitting and receiving signals in a full-duplex mode.

LTE is designed primarily for full-duplex operation in paired spectrum.Specifically, LTE employs a first frequency band for downlink and asecond frequency band for uplink. This is referred to as frequencydivision duplexing (FDD). In contrast, WiMAX operates in a half duplexmode in unpaired spectrum, where information is transmitted in onedirection at a time. This is typically referred to as time-divisionduplexing (TDD). As used herein, with respect to Step 302, full-duplexrefers to simultaneously transmitting and receiving in the samefrequency band.

The full-duplex mode may be used to provide for channel estimation andcovert data communications. During channel estimation, the transmittedsignals comprise known symbols and/or pilots. During data transmission,the signals comprise a known reference signal imparted with synthesizedchannel distortions that convey information. When both nodes 101 and 102simultaneously transmit in the same band, the SNR at an unintendedreceiver is reduced, making it difficult for the unintended receiver toaccurately estimate its channel or differentiate between natural andsynthesized channel distortions in the transmitted signals.

When two radio signals are received by an antenna, the voltage generatedat the antenna is the sum of the two signals. That voltage isdown-converted to a baseband signal and scaled such that the combinedsignal occupies a predetermined voltage range to ensure that the fulldynamic range of the receiver's ADC is used. This provides the bestpossible representation of the analog signal in the digital domain. Ifone signal is much smaller than the other signal, then it is representedby fewer bits than if the smaller signal arrived at the ADC by itself.

The transmitted signal can cause self-interference at the receivingantenna that exceeds the desired received signal of interest by as muchas 100 dB. With such a large difference in the signal powers, the finiteresolution of the ADC is the main bottleneck in enabling full-duplexcommunications. To achieve full-duplex communications over reasonabledistances, it is necessary to suppress the self-interference before itreaches the ADC.

FIG. 8 is a flow diagram of a method according to one aspect of theinvention that employs a combination of antenna cancellation 801, analogcancellation 802, and digital cancellation 803 to achieve full-duplexoperation. The isolation (in dB) achieved by employing multiple stagesof cancelation is additive.

In one aspect of the invention, separate transmit and receive antennasare employed. The antennas may be positioned in close proximity to eachother such that their channels are highly correlated. For example, theseparation between the antennas may be less than one wavelength. Thisclose proximity ensures that the channel measured by Node 1 issubstantially identical to the channel measured by Node 2. However, suchclose proximity causes substantial coupling (i.e., self-interference)between the transmitter and the receiver when operating simultaneouslyin the same band.

In one aspect of the invention, antenna cancellation may compriseproviding for passive interference suppression wherein the antennas arepositioned to reduce self-interference. For example, mobile devicestypically employ omni-directional antennas. However, such antennas arenot perfectly omni-directional, especially in the near field.Omni-directional antennas typically have small gain along the z-axis.Thus, the transmit and receive antennas may be mounted on top of eachother.

Passive interference suppression may comprise selecting a transceiver'santenna orientation and/or position to have less gain in the directionof another of the transceiver's antennas. In some aspects of theinvention, passive interference suppression may comprise shielding, suchas for attenuating antenna gain in a predetermined direction.

Antenna cancellation 801 typically comprises a passive cancellationmechanism, such as configuring a receive antenna system such that asubstantial portion of the transmit signal is cancelled at the receiver.Similarly, antenna cancellation may comprise configuring a transmitantenna system to produce a transmission null at the location of thereceive antenna system. In one aspect, antenna cancellation 801comprises combining received signals following RF receiver front-endprocessing 810. RF receiver front-end processing 810 may comprisefiltering and/or down-converting the received signals.

Antenna cancellation 801 may comprise an active cancellation mechanism,such as providing complex gains (e.g., frequency-dependent complexgains) to the received signals prior to combining for enhancinginterference cancellation. Similarly, antenna cancellation 801 maycomprise providing complex gains (e.g., frequency-dependent complexgains) to the transmitted signals in the transmitter's front-endprocessing step 820, such as to enhance cancellation of the node'stransmitted signal in its receiver. In one aspect of the invention,antenna cancellation comprises both transmit-side and receive-sideantenna cancellation for enhancing isolation.

Analog cancellation 802 is an active cancellation mechanism thatdelivers a canceling signal via an alternative radio chain to thereceiver. For example, a cancellation signal is synthesized from thetransmit signal, such as in the transmitter front-end processing step820, and combined with the received signal output from antennacancellation 801. Analog cancellation may be performed at the transmitfrequency (e.g., RF), an intermediate frequency, or at baseband.

Digital cancellation 803 is an active baseband cancellation mechanismthat synthesizes a cancellation signal from knowledge of the interferingtransmit signal. For example, digital processing 811 in the receiver maycomprise down-converting and digitizing the cancelled signal producedfrom analog cancellation 802 to produce a received digital signal. Thetransmitter's digital processing 811 synthesizes a digital cancellationsignal, which is combined with the received digital signal in thedigital cancellation step 803 to produce an interference-cancelleddigital signal.

FIG. 9 is a block diagram of a full-duplex transceiver configured inaccordance with an aspect of the invention to employ a combination ofantenna cancellation, analog cancellation, and digital cancellation.

A transmitter antenna system comprises transmit antennas 901 and 902,and a receiver antenna system comprises receiver antennas 951 and 952.The antennas 901, 902, 951, and 952 may be positioned in close proximityto each other such that the transmit and receive channels are highlycorrelated.

The transceiver depicted in FIG. 9 employs both transmit-side andreceive-side antenna cancellation. The antennas 951 and 952 of thereceive antenna system are positioned in the transmit antenna system'sbeam-pattern null 950, and the antennas 901 and 902 of the transmitantennas system are positioned in the receive antenna system'sbeam-pattern null 900. Specifically, the beam pattern null zone 950 ofthe transmit antennas 901 and 902 is a location where transmissions fromthe transmit antennas 901 and 902 are substantially cancelled.Similarly, signals produced by transmission sources located in thebeam-patter null 900 of the receive antennas 951 and 952 aresubstantially cancelled in the receive antenna system. Such antennaplacement schemes are well-known in the art, such as described in U.S.Pat. No. 6,211,671, which is hereby incorporated by reference. Forexample, the '671 patent shows antenna arrangements that provide forsubstantially identical, but out-of-phase transmit signals, that combinedestructively to cancel at the receiver. Similarly, the '671 patentshows receiver configurations for cancelling transmit leakage.Furthermore, the '671 patent shows analog signal cancellation.

Each transmit antenna 901 and 902 comprises a power amplifier, 911 and912, respectively, for amplifying up-converted transmit signals. One ofthe transmit branches may comprise a phase shifter 911 for providing a πphase shift to one of the transmit signals. A gain control, such as gaincontrol 910, may be provided to at least one of the transmit branches.The gain control 910 may provide for positioning the beam pattern nullzone 950 to effect transmit antenna cancellation.

Gain control and phase shifting may be provided to the transmit signalbefore or after up-conversion (not shown). While not shown, anup-converter may be provided in the splitter 905 or in the signal pathpreceding the power amplifiers 921 and 922 (but following the gaincontrol 910).

A signal is said to be nulled, or cancelled, when two copies of thesignal combine π out of phase to cancel each other, thereby reducing thereceived signal strength to or below the noise floor. The relative phasebetween the two signals may be controlled by introducing a phase offsetto one of the signals and/or by varying the relative distance betweenthe transmit antennas with respect to the receive antenna. Thus,positioning one antenna system in the beam-pattern null of anotherantenna system may comprise a combination of physically positioning oneor more of the antennas and providing complex gains to one or moreantenna signals, such as to select or adjust the beam pattern.

With a symmetric placement of a pair of transmit antennas relative to areceive antenna, a balanced to unbalanced transformer element or anyother type of a π-phase shifter inside one of the transmitters may beused to invert one of the transmit signals. Alternatively, suchcancellation may be achieved when there is a half-wavelength differencein the separations between the receiver antenna and the two transmitantennas. However, this approach by itself is only effective fornarrowband interference cancellation, as the required separations varywith respect to frequency. Thus, as the '671 patent notes, in multiband(e.g., OFDM) or wideband transmissions, frequency-dependent complexweights may be provided to the antenna signals in either or both thetransmit and receive antenna systems to enhance cancellation. Forexample, a different set of complex weights may be employed forweighting each subcarrier in an OFDM signal. Thus, adapting the complexgains provided to the transmit and/or received signals may be part ofthe antenna-cancellation scheme.

The receive antenna system comprises a gain control 954 and a phaseshifter 953 on one of the pair of receive branches. A combiner 955combines weighted, phase-shifted signals received from antenna 951 withsignals received from antenna 952 to provide receiver antennacancellation. Receive antenna cancellation is performed with thereceived analog signals at RF, an intermediate frequency, or atbaseband. In some aspects of the invention, the combiner 955 may performany necessary scaling and/or inversion of its input signals to producean interference-cancelled received signal.

In the transceiver depicted in FIG. 9, receiver antenna cancellation isperformed at RF. In accordance with one aspect of the invention,providing for passive interference suppression and/or antennacancellation eliminates the need for a power amplifier in an analogcanceller's RF chain. The analog canceller comprises an analog signalsynthesizer 940, which receives a digital transmit signal from abaseband processor 930. The analog signal synthesizer 940 generates ananalog cancellation signal, which is a replica of the transmit signal. Acombiner 960 combines the analog cancellation signal with theinterference-cancelled received signal produced by combiner 955. Thecombiner 960 may perform any necessary scaling and/or inversion of itsinput signals to produce an interference-cancelled analog signal.

The interference-cancelled analog signal is amplified by a low-noiseamplifier (LNA) 961 and down-converted by a down-converter 962 (e.g., amixer or a direct-conversion sampler), if necessary. An ADC 963 convertsthe down-converted analog signal to a digital receive signal. A digitalcanceller 965 synthesizes a digital cancellation signal from a digitaltransmit signal received from the baseband processor 930 and cancels thetransmit signal leakage in the digital receive signal to produce aninterference-cancelled digital signal. The interference-cancelleddigital signal may undergo further baseband processing, such as symbolestimation performed by a symbol estimator 969.

Step 303: Adaptive MIMO

Physical-layer security includes waveform security and channel security.Waveform security usually involves hiding transmitted information in thebackground noise. For example, ultra-wideband transmissions aretypically below the noise floor of narrowband receivers. Spread-spectrummodulation, such as direct sequence spread spectrum, spreads anarrowband information signal over a wide bandwidth. The privacy of suchdirect-sequence modulation depends on the security of the spreadingcode, which is typically a pseudo-random code.

Channel security typically comprises ensuring that any unintendedreceiver has a much lower SNR than intended transceivers. For example,beamforming techniques that deliver a highly focused transmission to adesired transceiver reduce the probability that the transmission can bedetected and intercepted by an unintended receiver. Beamforming canfurther reduce the SNR of signals received by unintended receivers viaspatially selective jamming. Short-range, low-power transmissions alsoreduce the probability of interception by unintended receivers.

In one aspect of the invention, a cooperative-MIMO system is employedfor spatially focusing transmission power to an intended receiver. Atransmitting node selects a predetermined number of local nodes tofunction as a cooperative antenna array. The selection may comprisedetermining the trustworthiness of the nodes, which may be performed viaany combination of automated processes and user inputs. For example, anauthentication procedure may be performed for authenticating eachcandidate node prior to selection. The trusted nodes may comprise apredetermined set of transceivers, such as transceivers operated by agroup of first responders or a team of military personnel.Authentication may comprise requiring a user input from the operator ofeach candidate transceiver to ensure that none of the selected nodeshave been compromised. The number of selected nodes may be determinedfrom any combination of parameters, including the number of unintendedreceivers within communication range, the number of spatial dimensionsemployed by each unintended receiver, the estimated SNR of the linkbetween the transmitting node and the intended receiving node, and thenumber of antennas employed by the intended receiver.

Determining the number of unintended receivers may comprise detectingtheir transmissions, identifying each node, and tracking each node. Oneaspect of the invention involves nodes snooping on headers of allongoing transmissions within radio range in order to identify and trackunintended receivers. Due to the close proximity of the cooperatingnodes, key distribution between the cooperating nodes can befacilitated, ensuring the security of the keys used to encrypt theirlocal communications. For local-group encryption in a cooperative-MIMOsystem, a shared secret key generated from channel estimates can be usedas a key sequence in a one-time pad, thus achieving virtually perfectsecrecy of the locally transmitted data.

Cooperative beamforming is performed by the selected nodes to improvethe power ratio γ of transmissions received by the intended receiverversus transmissions received by one or more unintended receivers. Forexample, in an exemplary multiple-input, single output (MISO) case, thesignal received by the desired receiver is y=h_(a) ^(T)wx+n_(a), and thesignal received by the undesired receiver is y_(u)=h_(u) ^(T)wx+n_(u),where h_(a) and h_(u) are the channel vectors, w is a vector ofbeamforming weights, and n_(a) and n_(u) additive white Gaussian noiseterms with a variance of σ_(i) ². The power ratio is expressed by

$\gamma = {\frac{{{h_{a}^{T}{wx}}}^{2}}{{{h_{u}^{T}{wx}}}^{2}} = \frac{w^{\dagger}h_{a}^{*}h_{a}^{T}w}{w^{\dagger}h_{u}^{*}h_{u}^{T}w}}$

The weight vector is selected to focus the transmitted signal at theintended receiver. If the power ratio γ is greater than one, thetransmit power can be adjusted to allow the intended receiver to decodethe transmission while providing any unintended receiver(s) with a lowSNR that prevents reliable detection.

An eigen-beamforming technique may be employed for selecting w. Forexample, in the MISO system, there is only one non-zero singular valuefor the channel. The transmit beamforming weight w is chosen to be thesingular vector corresponding to the non-zero singular value of thechannel h_(a). Even without precise knowledge of the channels ofunintended receivers, eigen-beamforming increases the power ratio γ bymaximizing transmission power received at the intended receiver. If thechannel h_(u) is known, the weight vector w may be chosen from a vectorin the null space of h_(u)*h_(u) ^(T)

In one aspect of the invention, the transmitter determines the minimumSNR necessary for the intended receiver to decode the transmission andadjusts the transmit power to reduce the likelihood that the unintendedreceiver(s) could decode the transmission.

FIG. 10A depicts an aspect of the invention wherein the first node 101employs beamforming to reduce the SNR of its peer-to-peer transmitsignal received by the third-party network 103. For example, aninformation-bearing transmit signal x_(n) comprises a first beam patterns₁ having substantially higher gain in a spatial subchannel used tocommunicate with the second node 102 compared to a spatial channelemployed for communicating with the third-party network 103. Similarly,beamforming may be employed to reduce the SNR of signals received bypotential eavesdroppers.

In one aspect of the invention, either or both of the nodes may transmita decoy signal, such as during peer-to-peer transmission of aninformation-bearing signal. The decoy signal may be transmitted to thethird-party network, such as to hold the assigned channel(s) employed bythe peer-to-peer link. In some aspects, the decoy signal may be intendedfor reducing the SNR of signals received by unintended receivers.

In an aspect of the invention, the first node may transmit a referencesignal used by the second node for decoding the information-bearingsignal transmitted via the peer-to-peer link. For example, the referencesignal may be transmitted via the third-party network, which the secondnode receives in a different radio channel than the radio channel(s)employed in the peer-to-peer link. Alternatively, the reference signalmay be transmitted in the peer-to-peer link, but via an alternativeradio channel than the radio channel(s) employed for transmitting theinformation-bearing signal.

FIG. 10B depicts an aspect of the invention wherein the first nodeemploys beamforming to null a decoy signal or reference signal in aspatial subchannel used for communicating with the second node. Forexample, a decoy interference signal x_(d) comprises a second beampattern s₂ having a high gain in a spatial subchannel used forcommunicating with the third-party network and substantially zero gainin a spatial channel employed by the peer-to-peer link. Similarly, thesecond beam pattern s₂ may be configured to provide high-gaintransmissions to unintended receivers.

In an aspect of the invention, a first transceiver transmits a signal toa second transceiver that is in the same frequency band as a signaltransmitted by a third-party transmitter. The first transceiver's signalcomprises known symbols, which are used for channel estimation. Thissignal also comprises a known reference signal imparted with synthesizedchannel distortions, which are information-bearing. Interference due tothe third-party's transmission makes it difficult for an unintendedreceiver to differentiate between the natural and synthesized channeldistortions in the first transceiver's transmission, especially if theundistorted reference signal employed by the first transceiver is thesame signal transmitted by the third-party transmitter. The secondtransceiver employs the channel information of the channel between thefirst and second transceivers (which is not known by any unintendedreceiver) and uses the third-party signal as a reference signal todecode the first transceiver's transmission.

In another aspect of the invention, a first transceiver transmits afirst signal to a second transceiver and a second signal to at least oneunintended receiver. The first signal comprises known symbols used bythe second transceiver for channel estimation. The first signal alsocomprises a reference signal imparted with synthesized channeldistortions that are data-bearing. The first transceiver employs antennaarray processing (such as via Cooperative-MIMO) to reduce the power ofthe second transmission at the second transceiver. For example, thefirst transceiver may employ beam-pattern null steering or some otherpre-coding that cancels the second transmission at the secondtransceiver. The second transmission may comprise a decoy signal, suchas the reference signal imparted with synthesized random channeldistortions.

FIG. 11 is a flow diagram of a method employing adaptation of acooperative-MIMO system according to aspects of the present invention.Channel measurements are collected 201, an antenna array size (i.e., thedesired number of antenna elements) is determined 202, the arrayelements are selected 203, and a secret key is generated 204.

Various parameters may be employed for determining the array size 202.For example, an SNR may be measured or calculated 211. Antenna arrayparameters (e.g., the number of antenna elements and the degree ofcorrelation between the elements) affect the achievable key-generationrate. Selecting the number of antenna elements may be highly dependenton the SNR of the channel. At low SNR, small antenna arrays outperformlarger arrays for key generation due to the lower BER. However, forhigher SNR, it may become advantageous to increase the array size, sinceincreasing the array size can improve key-generation efficiency athigher SNR. Antenna selection (i.e., selecting which antennas comprisethe array) may be performed to improve the SNR of the measured signals.

Unintended receivers that are potential eavesdroppers may be detected212 and/or tracked. The number and locations of unintended receivers maybe a factor in determining the array size 202. The desired key rate maybe calculated 213 based on the amount of data that needs to betransmitted securely. Since larger arrays typically enable a highernumber of uncorrelated channel measurements to be collected 201, higherkey rates may be achieved. The array size may depend on the availabilityof cooperating nodes 214 that can be employed in a cooperative-MIMOconfiguration.

Once the array size is determined 202, array elements may be selected(or de-selected) 203. If no changes are required for the array, then thesecret key is generated 204 using the most recent set of channelmeasurements. When array elements are selected (or de-selected) 203, newchannel measurements may be collected 201, such as to compare the SNR ofthe current array configuration to previous configurations. In someaspects of the invention, secret key generation 204 may be performedfollowing array selection 203, even while the array configuration isbeing updated.

Step 304: Channel Measurement

In the standard multipath channel model, a received signal y(t) isexpressed as a convolution of a transmitted signal x(t) with a channelimpulse response h(t, τ) of the time-varying multipath channel, where τexpresses the multipath delays for a fixed value of t:y(t)=x(t)

h(t,τ)

Since the received signal y(t) comprises a series of attenuated,time-delayed, phase-shifted replicas of the transmitted signal x(t), thebaseband impulse response h(t, τ) of the multipath channel can beexpressed as

${h\left( {t,\tau} \right)} = {\sum\limits_{i = 0}^{N - 1}\;{{a_{i}\left( {t,\tau} \right)}e^{i{({{2\pi\; f_{c}{\tau_{i}{(t)}}} + {\phi_{i}{({t,\tau})}}})}}{\delta\left( {\tau - {\tau_{i}(t)}} \right)}}}$where a_(i)(t, τ) is the real amplitude and τ_(i)(t) is the path delayof an i^(th) multipath component. The phase term φ_(i)(t, τ) representsthe phase shift of the i^(th) multipath component. Nis the total numberof multipath bins, and the delta function δ( ) is a unit impulsefunction that designates which bins (delays τ_(i)) have multipathcomponents.

FIG. 12A is a block diagram of an OFDM transmitter configured inaccordance with certain aspects of the invention. In an OFDMtransmission, some of the OFDM sub-carriers may comprise pilot signalsused for channel measurements (i.e., the equalizer gain and phase shiftfor each sub-carrier). Pilot signals and training symbols (e.g.,preambles) may also be used for time synchronization to avoidinter-symbol interference and frequency synchronization to avoidinter-carrier interference caused by Doppler shift.

Data bits are grouped and mapped to symbols by a symbol mapper 1201. Apilot-insertion module 1202 periodically inserts pilots on all the OFDMsubcarriers (i.e., in all the OFDM frequency bins) at predetermined timeintervals, or the pilots are continuously provided to predeterminedsubcarriers. An IDFT 1203 transforms the data sequence X(k) of length Ninto a time-domain signal, x(n): x(n)=IDFT{X(k)} n=0, . . . , N−1.

A cyclic-prefix pre-pender 1204 selects a guard time that is larger thanthe expected delay spread, and pre-pends a cyclically extended part ofthe time-domain signal in order to eliminate inter-carrier interference(ICI). The resulting OFDM symbol is given as follows:

${x_{f}(n)} = \left\{ \begin{matrix}{{x\left( {N + n} \right)},} & {{n = {- N_{L}}},\ldots\mspace{14mu},{- 1}} \\{{x(n)},} & {{n = 0},\ldots\mspace{14mu},{N - 1}}\end{matrix} \right.$where N_(L) is the length of the guard interval.

The transmitted signal x_(j)(n) passes through a frequency-selectivetime-varying fading channel with additive noise. The received signal isgiven by:y _(f)(n)=x _(f)(n)

h(n)+w(n)where w(n) is AWGN and h(n) is the channel impulse response.

FIG. 12B is a block diagram of an OFDM receiver configured in accordancewith certain aspects of the invention. The received signal is digitizedin an ADC 1211, filtered by a low-pass filter 1212, and the cyclicprefix is removed in block 1213 before DFT processing in DFT block 1214.The output of the DFT 1214 is given by Y(k)=DFT{y(n)}, k=0, . . . , N−1

${Y(k)} = {\frac{1}{N}{\sum\limits_{n = 0}^{N - 1}\;{{y(n)}e^{- {i{({2\pi\;{{kn}/N}})}}}}}}$

Y(k)=X(k)H(k)+W(k), where W(k)=DFT{w(n)}.

Following the DFT 1214, a channel estimator 1215 extracts the pilotsignals, and the estimated channel H_(e)(k) for the data sub-channels iscalculated. Then the transmitted symbols are estimated, such as by:{circumflex over (X)}(k)=(H_(e)(k))⁻¹Y(k). A symbol demapper 1216estimates the transmitted binary data from the symbol estimates.

Channel estimation at pilot frequencies may be based on LS, MMSE, or LMSmethods. MMSE has been shown to perform much better than LS, and thecomplexity of MMSE can be reduced by deriving an optimal low-rankestimator with singular-value decomposition. In some aspects of theinvention, channel estimation based on a block pilot arrangement isperformed by transmitting pilots in every sub-channel and using thechannel estimation for preceding and/or following symbols. For a slowfading channel, the channel estimation inside the block can be updatedusing decision feedback equalization at each sub-carrier.

In a fast-fading channel, a comb-type channel estimation (wherein thetransmitted signal comprises dedicated pilot tones) may be employed.Channel estimation for a comb-type pilot arrangement may furthercomprise interpolation. Channel interpolation may include linearinterpolation, second-order interpolation, low-pass interpolation,spline-cubic interpolation, or time-domain interpolation. Time-domaininterpolation is typically performed using an inverse discrete Fouriertransform (IDFT), zero padding, and returning to the frequency domainvia a discrete Fourier transform (DFT).

In one aspect of the invention, the time intervals employed for blockpilots are selected based on a function of the secret key derived fromchannel estimates. In another aspect of the invention, at least thepilot subcarriers for comb-type channel estimation are selected based ona function of the secret key derived from channel estimates.

Step 305: Generate Secret Key

Aspects of the invention are not limited to any particular techniquesfor selecting which channel measurements are used to generate a secretkey. In some aspects, keys may be generated by discretizing extractedcoefficients of the multipath components or quantizing the channelphases. For example, the phases of the reciprocal channel may be used torandomly rotate the phase of each transmitted data symbol, whereinencryption comprises employing one key symbol per data symbol. Oneaspect of the invention exploits amplitude level crossing of themeasured channel coefficients to robustly generate keys with lowmismatch probability. Some aspects of the invention provide forseparately quantizing real and imaginary parts of the channelcoefficients, since the real and imaginary parts are presumed to beindependent of each other.

FIG. 13 is a flow diagram depicting a method for generating a secret keyfrom channel measurements in accordance with aspects of the invention.Correlated measurements are combined 1301 to enhance the SNR of thevalues to be quantized 1302. Error correction 1303 provides forcorrecting key mismatches between the nodes 101 and 102. Optionally,privacy amplification 1304 may be performed.

In practice, MIMO subchannels may exhibit some degree of correlation dueto a poor scattering environment or closely spaced antennas.Consequently, the achievable capacity in actual propagation environmentsis typically lower than the theoretical capacity. According to someaspects of the invention, samples from correlated channels are combined1301 or averaged to provide combined measurements having a higher SNR.

Quantization 1302 of the observed channel parameters can lead tomeasurement disagreements between two nodes establishing a channel-basedkey, thus resulting in key mismatch. In practical systems, differencesin the way the channel is measured can result in discrepancies betweenthe channel estimates generated by the nodes 101 and 102. Other factors,such as noise and external interference in the radio channel can alsolead to differences in channel estimates, and thus, key disagreements.

In some aspects of the invention, quantizing the channel measurementsmay comprise calibrating the transceivers 1311 to mitigate the effectsof such discrepancies and differences, exchanging quantization and(optionally) other measurement information 1312, determiningquantization levels 1313, adapting quantization decision boundaries1314, and discarding some of the measurements 1315.

In some aspects of the invention, quantization algorithms are employedthat transmit information 1312 to improve key agreement withoutrevealing information about the key to an unintended receiver. Thenumber of quantization levels is generally selected 1313 based on thechannel SNR and the desired encryption key generation rate. In someinstances, the key rate may be constrained by the available SNR. In someaspects of the invention, adaptive (i.e., dynamic) quantization may beemployed. For example, each node 101 and 102 may determine thequantization levels 1313 in an iterative method that comprisesexchanging quantization information 1314. Typically, quantizationinformation, such as region boundaries and number of quantization levelsis shared via a public channel between the transceivers. However,quantization parameters may be determined directly from the channelmeasurements as if the quantization levels are selected based on the SNRof the channel vectors.

Furthermore, each node 101 and 102 may adapt its decision boundaries1314 during a training phase to reduce channel-estimation errors,thereby improving key agreement. Such boundary adaptations 1314 may bemade in response to measurement information transmitted by the othernode (101 or 102) in step 1312. In some aspects, measurements nearquantization boundaries may be discarded 1315.

In one aspect of the invention, channel quantization with a guard bandmay be employed. To exploit both amplitude and phase, the space ofobservable complex channels is divided into equally probablequantization sectors, and each sector is assigned a unique bit pattern.As the nodes 101 and 102 observe the channel at predetermined sampletimes, the bits in the corresponding sector are added to the key.

A guard band is used to reduce the probability of key mismatch bydiscarding channel measurements 1315 observed in the guard-band region.In a one-way handshake, Node 1 transmits a guard band indicator bit toNode 2 over a public channel indicating an observation of the channelinside or outside the guard band. Both nodes discard channelobservations inside the guard band. In a two-way handshake, both nodesexchange guard band indicator bits and discard the measurement if eithernode indicates that the observation is inside the guard band. Increasingthe guard band reduces the key-generation rate, but also reduces thesymbol-mismatch rate.

Some aspects of the invention may employ error correction strategies1303, such as LDPC coding to correct for key mismatch. The SNR requiredfor key generation can be reduced by using LDPC codes in conjunctionwith other BER reduction techniques. LDPC codes allow MIMO to beexploited with significantly reduced SNR at the expense of a reductionin the key-generation rate by a factor of approximately two(corresponding to the rate 1/2 code).

LDPC coding provides a coding gain at low SNR, thus allowing a targetkey-generation rate to be achieved at lower SNRs. However, at higherSNRs, higher code rates (i.e., fewer redundant or parity-check bits) maybe used. At a sufficiently high SNR, the coding gain is no longernecessary, so LDPC coding 1303 can be eliminated. When the SNR is high,channel estimation errors are small, and the errors tend to be primarilyin estimates near quantization boundaries. Adaptive quantization (e.g.,1313 and 1314) and discarding estimates near quantization boundaries1315 may be the principal error-mitigation techniques employed.

A method according to one aspect of the invention comprises each node101 and 102 calculating the degree of channel correlation (e.g., delayspread) in its MIMO-OFDM channel measurements and selecting apredetermined set of uncorrelated subcarrier frequency/sub-space pairs.The nodes 101 and 102 may employ cooperative-MIMO to increase the numberof independent spatial channels, thereby increasing the key-generationrate.

The nodes 101 and 102 identify correlated measurements (e.g., correlatedchannels) and combine the correlated measurements 1301 to producecombined measurements with increased SNR.

Channel measurements may be collected while performing full-duplexsimultaneous pilot transmission and channel sounding in which thetransmit signal is cancelled at the receiver.

The first node 101 estimates the full covariance from the combinedestimates of its channel vector h_(a) and decorrelates h_(a) using theeigenvectors U_(a). For example, in a MIMO system with N_(t) transmitantennas and N_(r) receive antennas, the channel from the transmitter tothe receiver is an N_(r)×N_(t) channel matrix, H, and the reverse linkfrom the receiver to the transmitter is an N_(t)×N_(r) channel matrix,H, where H=H ^(T). These matrices are represented by an N_(r)N_(t)×1vector, h=vec{H}=vec{H ^(T)}. Node 101 computes the covarianceR=E{hh^(T)} and the eigen-decomposition, R=UVU^(T), where A is thediagonal matrix of non-negative real eigenvalues and U is the unitarymatrix of eigenvectors. Then node 101 constructs the decorrelatedchannel vector ĥ=U^(T)h.

Node 101 generates a key using a coefficient quantization algorithm withflexible quantization levels determined by the SNR for each element ofthe channel vector. Node 101 also determines the quantization-map bitsand forms syndromes of the key's binary sequence by multiplying thebinary sequence with a parity-check matrix of the LDPC codes.

The eigenvectors, quantization levels, quantization-map bits, andsyndromes of the binary bits are transmitted through the publicinsecure, but authenticated channel from Node 101 to Node 102. If aneavesdropper is an active attacker, a secure channel may be provided toprotect data integrity of the transmissions. For example, the nodes 101and 102 may share an initial key prior to transmission.

Node 102 performs a decorrelation of the received eigenvectors, and,based on the quantization regions determined by the quantization-mapbits and quantization levels, generates a key binary sequence using thesame coefficient quantization algorithm. An estimate of Node 101'sbinary sequence is obtained with the help of the syndromes and Node102's binary sequence. Some aspects of the invention may employ softdecisions or mixed decision processing (i.e., a combination of soft andhard decisions) in the detection of the binary sequence using LDPC codesIf the syndromes of the bit sequences are transmitted over the publicchannel, privacy amplification may be performed on the binary sequencesusing a universal class of Hash functions for producing the final secretkey.

Step 306: Coordinate Channel Updates

In one aspect of the invention, the nodes 101 and 102 estimate theirpeer-to-peer reciprocal channel using the same training sequences and/orpilot tones that the third-party network 103 uses for channelestimation. In another aspect of the invention, the nodes 101 and 102may employ the data-payload portion of the third-party network's 103frame format for transmitting known symbols, which are used forestimating the peer-to-peer channel. In either aspect, a selection oftraining symbols (e.g., subcarriers selected for pilot tones and/or timeslots during which known symbols are transmitted) may be based on secretkey. Since both nodes 101 and 102 generate the same secret key, that keymay be used to select pilot tones and/or data sequences at atransmitting node, and the key may be used at a receiving node forselecting subcarrier frequency bins and/or time slots in the receivedsignal for measuring the peer-to-peer channel.

FIG. 14A is a block diagram of a transmitter configured for transmittingknown symbols in time slots and/or subcarrier frequency bins determinedfrom a function of the secret key. A key generator 1401 supplies thesecret key to at least one of a time-slot selector 1402 and afrequency-bin selector 1403. The time-slot selector 1402 selects asequence of time slots based on a predetermined function of the secretkey, wherein the predetermined function is employed by both nodes 101and 102. The frequency-bin selector 1403 selects one or more frequencybins based on a predetermined function of the secret key, wherein thepredetermined function is employed by both nodes 101 and 102.

In some aspects of the invention, the frequency-bin selector 1403 maygenerate a combination of time slots and frequency bins for transmittingpilot tones, such as in a frequency-hopped OFDM system or a block-typeOFDM channel-estimation scheme. In a comb-type channel-estimationscheme, the pilot tones may be varied with respect to time. Atraining-symbol generator 1404 generates the known symbols and maps thesymbols onto the frequency bins of an IDFT 1405 in accordance with thefrequency bin selections and/or time-slot selections.

FIG. 14B is a block diagram of a receiver configured for performingchannel estimation in accordance with an aspect of the invention. Areceived baseband signal is separated into its frequency components by aDFT 1415. Frequency bins comprising known training symbols or pilottones are selected for processing by a channel estimator 1414. Thechannel estimator 1414 is responsive to at least one of frequency-bininformation and timing information received by at least one of afrequency bin selector 1413 and a time-slot selector 1412, respectively.A key generator 1411 supplies a secret key (which is preferablyidentical to the secret key generated by key generator 1401) to thefrequency-bin selector 1413 and the time-slot selector 1412. Thefrequency-bin selector 1413 operates in a similar manner as thefrequency-bin selector 1403, and the time-slot selector 1412 operates ina similar manner as the time-slot selector 1402.

Step 307: Adapt Data to Key Rate

According to one aspect of the invention, when the data to betransmitted exceeds the maximum key rate, data to be transmitted may bedivided into a sensitive data portion and a non-sensitive data portionin a data-sanitization step. For example, the data may be separated suchthat the data rate of the sensitive portion is less than or equal to themaximum key rate. An encryption process providing a high security levelis applied to the sensitive data, and a relatively less secure processmay be applied to the non-sensitive data prior to transmission.

As used herein, data sanitization comprises the process of separatingsensitive information from a document or other medium. When dealing withclassified information, sanitization can reduce the document'sclassification level, possibly yielding an unclassified document.Data-sanitization may comprise redaction, which generally refers to theediting or blacking out of text in a document, such as to allowselective disclosure of information in the document while keeping otherparts of the document secret. With respect to aspects of the invention,such applications of data sanitization comprise separating essential orsensitive information from data, documents, images, audio, video, and/orother media to generate a smaller data portion for encryption, such as ahighly secure encryption algorithm constrained by a low key generationrate.

Step 308: Select Reference Signal

FIG. 15 is a flow diagram depicting a communication method in accordancewith an aspect of the invention. The nodes 101 and 102 agree upon acommon reference signal 1501. For example, a signal broadcast by a thirdnode (such as a base station, access point, or another client in athird-party network 103) is received by the first node 101 andrebroadcast with information-bearing channel distortions to the secondnode 102. The selected reference signal may comprise a network controlchannel or portions of a broadcast signal, such as pilot sequences inthe downlink transmitted by a base station.

Each node 101 and 102 performs channel estimation 1502 of its reciprocalpeer-to-peer channel and its channel with the third-party network. Thisenables the second node 102 to employ at least a first spatial channelfor receiving transmissions from the first node 101 and at least asecond spatial channel for receiving transmissions from the third-partynetwork. Thus, at least one of the first node 101 and the second node102 may employ spatial processing for providing a clean reference signalto the second node 102 via a separate channel 1503 from the channelemployed in the peer-to-peer link. The first node 101 transmits aninformation-bearing reference signal 1504 to the second node 102 via thepeer-to-peer link. Upon separating the received clean reference from theinformation-bearing reference, the second node 102 decodes theinformation-bearing reference 1505.

FIG. 16A depicts an aspect of the invention wherein a third-partytransmission d is employed as the selected reference signal. The firstnode 101 receives the transmission d and transmits an estimated version{circumflex over (d)}₁ imparted with synthesized channel distortions W.The information-bearing reference signal is received at the second node102 after being distorted by the reciprocal peer-to-peer channel denotedby H. The second node 102 may employ spatial processing to separate itsreceived third-party transmission d from the receivedinformation-bearing reference signal.

The node 102 employs its channel estimate of H and its estimate of thereference d to decode the received information-bearing reference.

FIG. 16B depicts an aspect of the invention wherein the first node 101transmits a clean reference signal d via a third-party network 103 and adistorted reference signal Wd with information-bearing synthesizedchannel distortions W via a peer-to-peer channel to the second node 102.The second node generates an estimated reference signal {circumflex over(d)} from the signal received from the third-party network 103, which,along with the channel estimate of H, is used to decode the receivedinformation-bearing reference signal.

In one aspect of the invention, node 101 employs its third-partyassigned uplink channel to transmit both the clean reference d and theinformation-bearing reference Wd. Node 101 may employ a first spatialchannel for transmitting the clean reference d and a second spatialchannel for transmitting the information-bearing reference Wd, whereinthe first spatial channel is nulled at the second node 102 and thesecond spatial channel is nulled at the third-party network'stransceiver 103. The second node employs its third-party assigneddownlink channel to receive the clean reference d.

In another aspect of the invention, the first node 101 employs itsthird-party assigned uplink channel to transmit the clean reference dand transmits the information-bearing reference Wd in an alternativechannel, such as the second node's 102 third-party assigned uplinkand/or downlink channel.

In another aspect of the invention, as depicted in FIG. 17A, the firstnode 101 employs a first peer-to-peer channel for transmitting a cleanreference signal d and a second peer-to-peer channel for transmitting aninformation-bearing reference signal Wd 1504. The second node 102employs spatial processing for separating the received clean referencefrom the information-bearing reference, and employs its estimate of theclean reference d and its estimate of the reciprocal channel H to decodethe received information-bearing reference.

In another aspect of the invention, which is depicted in FIG. 17B, thesecond node 102 generates a local version of the clean reference signald. For example, the reference signal d may be predetermined and known byboth nodes 101 and 102 prior to establishing the peer-to-peer link. Inone aspect of the invention, the locally generated clean referencesignal d may comprise a signal stored in memory in node 102.

Step 309: Synthesize Channel Distortions

In accordance with one aspect of the invention, FIG. 18 depicts a methodfor providing a communication link with LPI/LPD properties. Channelestimation 1801 is performed by each of at least a pair of nodes betweenwhich a covert communication link will be established. The channelestimates may comprise flat-fading channel estimates, channel impulseresponse estimates, subspace channel estimates, or any combinationthereof.

For each subcarrier frequency of a MIMO-OFDM system, the data sequenceis split into N_(T) sub-sequences that are transmitted simultaneouslyusing the same subcarrier frequency band. The resulting data rateincrease (i.e., the spatial multiplexing gain) can be up to a factor ofN_(T) if N_(R)≧N_(T). At the receiver, the sub-sequences are separatedby using an interference cancellation algorithm (e.g., linearzero-forcing (ZF), minimum-mean squared-error (MMSE), maximum-likelihood(ML), successive interference cancellation (SIC)).

For example, FIG. 19A illustrates a bidirectional channel estimationstep in which nodes 101 and 102 each transmit a known training sequenced₀ or pilot signal, which is used to estimate the channel H. Node 102generates an estimated channel matrix H₁₂=H+ΔH₁₂ at a predetermined timesample wherein H is the true N_(R)×N_(T) channel matrix for the forwardlink from Node 101 to Node 2. Node 101 generates an estimated channelmatrix H₂₁=H ^(T)+ΔH₂₁ at a predetermined time sample wherein H ^(T) isthe true N_(T)×N_(R) channel matrix for the reverse link from Node 102to Node 1. The matrices ΔH₁₂ and ΔH₂₁ represent channel estimationerrors, which are typically regarded as independent random variableshaving zero mean with a noise variance σ_(n) ².

Prior to channel estimation 1801, signaling parameters, such as thefrequency band and the number of antennas employed by each node 101 and102, are determined. For example, the nodes may employ a third-partynetwork, such as a cellular communications network, to assigncommunication channels to the nodes and reserve those channels for asession interval during which the nodes 101 and 102 establish a covertcommunication link. The nodes may employ any combination of theirassigned third-party uplink and downlink channels for the covertcommunication link. In one aspect of the invention, third-party channelassignments are shared between the nodes 101 and 102. In another aspectof the invention, third-party network identifiers are shared between thenodes such that each node 101 and 102 can listen for the other nodeschannel assignments.

In some aspects of the invention, the nodes 101 and 102 may employ athird-party broadcast channel as the channel for the covertcommunication link. Thus, prior to channel estimation, at least Node 102may perform spatial processing, such as to null (e.g., reduce itssensitivity to) a broadcast channel or downlink channel employed by thethird-party network 103.

Channel estimation 1801 may be performed periodically, such as whenknown training sequences are transmitted, and/or concurrently with datatransmissions, such as in OFDM signaling, which commonly employs pilottones. Channel estimates are typically updated as the channel changesand when system changes occur, such as when changes to channelassignments or the number of antennas occur.

In some aspects of the invention, channel estimation 1801 may beperformed in a full-duplex mode. For example, in one aspect of theinvention, Nodes 101 and 102 simultaneously transmit known signals inthe same frequency band while cancelling the self-interference. Inanother aspect of the invention, Nodes 101 and 102 employ frequencydivision duplexing (FDD) to transmit known signals in differentfrequency bands whose channels are highly correlated. In yet anotheraspect of the invention, Nodes 101 and 102 employ time divisionduplexing (TDD) for transmitting known symbols in different timeintervals during which the channels are highly correlated.

Node 101 synthesizes information-bearing channel distortions 1802, whichare imparted on a reference signal, which is transmitted to Node 1.Specifically, the information is disguised as channel distortions andmay take the form of multipath components, variations in the complexcoefficients in a flat fading channel, beam-pattern variations (such asin spatial multiplexed or MIMO signals), or any combination thereof.

In one aspect of the invention, the channel distortions are encodedusing the channel estimates calculated during channel estimation 1801.Synthesizing the channel distortions 1802 may comprise establishing apredetermined set of signaling parameters 421, selecting which signalingparameters to vary based on the channel estimates 422, varying theselected signal parameters with respect to an information signal 423,and, optionally, varying the remaining signal parameters with respect toa decoy signal.

In accordance with one aspect of the invention, the predetermined set ofsignaling parameters may comprise a set of subcarriers in an OFDMsignal, a set of subspaces in a MIMO signal, and/or a set of delays(such as determined with respect to a channel impulse response). Node101 selects the set of signal parameters with respect to the channelestimate H₂₁. For example, Node 101 employs some predetermined formulato select which subcarriers of the reference signal to distort based onits channel estimate H₂₁. Since Node 102 generates a channel estimateH₁₂ that is substantially identical to H₂₁, using the same predeterminedformula, Node 102 determines which signal parameters of the receivedreference signal may comprise information-bearing distortions. Node 101may generate a data sequence comprising error-correction coding, such asparity check bits, which it then uses to distort the selected signalparameters. By using error correction coding, any detection errors orkey disagreements between the Nodes 101 and 102 can be identified andpossibly corrected.

If Node 101 employs beamforming weights (e.g., a beam-forming matrix Wfor precoding data symbols d to produce a data signal vector, x=Wd), thereceived signal y at Node 102 is y=HWd. The effective channel matrix atNode 102 is H_(eff)=HW. If Node 102 does not have an accurate estimateof the channel matrix H, the beam-forming matrix W portion of H_(eff) isindistinguishable from the channel matrix H portion. Thus, aspects ofthe invention that corrupt an eavesdropper's channel estimates maysynthesize information-bearing beamforming weights that essentiallydisguise information within the random communication channel.

In the case of a quadratic MIMO system wherein the number of transmitantennas and the number of receive antennas are equal (N_(T)=N_(R)), thefirst node 101 constructs a precoded signal vector x′=W′d, wherein thedata signal vector d is multiplied by a precoding matrix W′. In oneaspect of the invention, the precoded signal vector x′ and the datasignal vector x=Wd are transmitted concurrently. Specifically, thetransmitted signal is x′+x=(W′+W)d. Thus, the precoding matrix W′ may beregarded as an additional beam-pattern feature added to the first node'sprimary transmit beam pattern (i.e., the beam pattern corresponding tothe transmission of signal x). The beam pattern of the W′ component maycomprise a beam pattern null in the direction of at least one unintendedreceiver. The coding of the information-bearing variations may be basedon the estimated channel between the transmitter and the receiver. Forexample, a combination of how the precoding matrix W′ varies and when itvaries may be based on channel estimation values.

In one aspect of the invention, the first node 101 employs theadditional beam-forming matrix W′, such as depicted in FIG. 19B, fortransmitting an uplink signal assigned by a third-party network 103. Theuplink signal is also received by the second node 102. If thetransmitted signal is x′+x=(W′+I)d, the uplink signal received by thesecond node 102 comprises y=HW′d+Hd+n. The second node 102 estimates thechannel matrix H when the first node 101 transmits known data signals(and/or pilots) in the reference signal d using only its primary beampattern.

The reference signal d is provided to the second node 1803. For example,the reference signal d may be a predetermined signal, such as a datasequence stored in memory at the second node or a locally generated datasequence that is a replica of the transmitted reference signal d. Insome aspects of the invention, the reference signal d is shared betweenthe nodes 101 and 102 prior to establishing a communication link. In oneaspect of the invention, the reference signal d is a broadcast signalreceived from the third-party network 103 or some other networkemploying a different communication channel assignment than the uplinkchannel assigned to node 101.

In accordance with one aspect of the invention, the reference signal dis transmitted by the first node 101 to the second node 102 via thethird-party network 103, such as depicted in FIG. 19B. The second node102 receives the reference signal from the third-party network on itsassigned downlink channel and processes the received signal to generatean estimated reference signal {circumflex over (d)}.

Using estimated values of H and {circumflex over (d)}, the second node102 synthesizes a cancellation signal 1804 which is subtracted from thereceived signal 1805. Thus, the term Hd can be cancelled 1805 from y,which yields an interference-cancelled signal, y′=HW′d.

The interference-cancelled signal y′ is equalized 1806. For example, inthe case of a zero-forcing detector, if H is full rank, then linearzero-forcing detection yields a post-processed received signal expressedby z′=H⁻¹y′=W′d+H⁻¹n. Equalization 1806 may comprise alternative typesof detection. For example, linear MMSE, ML, or SIC may be employed.

Decoding the information-bearing matrix W′ 1807 may comprise removingthe reference signal d. In one aspect of the invention, the data-bearingelements of W′ are identified by some predetermined function of theestimated channel matrix H. Thus, the channel estimates H₁₂ may beemployed for decoding W′.

FIG. 20 is a block diagram of a transmitter configured in accordancewith an aspect of the invention. A reference signal generator 2001generates a reference signal that is employed as a carrier signal forinformation-bearing distortions generated to resemble channeldistortions. A channel estimator 2002 provides estimates of acommunication channel between the transmitter and a receiving node (suchas the receiver shown in the block diagram of FIG. 21).

In some aspects of the invention, the channel estimates are used togenerate a secure key. A similar (and, preferably, identical) key isgenerated by the receiving node and is used by the receiving node todecode its received signals. The secure key may be employed in adistortion synthesizer 2003. For example, the secure key may be input toa distortion synthesizing function that selects a set of signalparameters to be modified with respect to a data signal from a datasource 2004.

In one aspect of the invention, the reference signal is an OFDM signal,and the distortion synthesizer 2003 selects which OFDM subcarriers willcomprise data-bearing distortions. The distortion synthesizer 2003 maygenerate complex weights that resemble flat fades on the selectedsubcarriers. In another aspect of the invention, the reference signal isa MIMO signal, and the distortion synthesizer 2003 selects combinationsof beam-forming coefficients. The beam-forming coefficients may beselected to enhance and/or suppress predetermined multipath componentsin accordance with a function of the channel estimates. In some aspectsof the invention, the distortion synthesizer 2003 selects a set of delayintervals that can be used to convey data. Thus, the distortionsynthesizer 2003 may synthesize time offsets that appear as multipathdelays to unintended receivers. In another aspect of the invention, thedistortion synthesizer may generate jitter and/or frequency offsets inthe reference signal in accordance with a combination of the secret keyand the data. In each of these cases, the information-bearingdistortions imparted to the reference signal are typically removed(e.g., equalized) by the front end of an unintended receiver.

A multiplier 2005 is depicted for imparting the distortions onto thereference signal. For example, an OFDM reference signal may bemultiplied by a vector of complex weights, or a MIMO reference signalmay be multiplied by an antenna array weighting matrix. In anotheraspect of the invention, an adding function (not shown) may be employedin place of the multiplier 2005 for adding distortions to the referencesignal. Additional aspects of the invention may employ alternativefunctions (not shown) for distorting the reference signal.

A multiplexer 2006 may be employed for multiplexing the distorted andundistorted reference signals. For example, the undistorted referencesignal may be transmitted at predetermined times to enable the receiverto perform channel estimation. Similarly, the undistorted signal may betransmitted on predetermined subcarriers to enable the receiver toperform channel estimation. Since the reference signal is predeterminedor otherwise known at the receiver, channel estimation does not need tobe confined to just OFDM pilot tones. Rather, at least some of the datasubchannels can be used for channel estimation. Similarly, at least someof the pilot subchannels may be provided with information-bearingdistortions. In other multiplexing schemes, the data payload of a framemay be employed for conveying training sequences, such as sequences thatmay be used to estimate the channel.

Using known channel state information at the transmitter and receiver, aMIMO system can employ singular value decomposition (SVD) beamforming toeffectively create parallel independent subchannels in space. Thesespatial subchannels typically possess different levels of SNR. Desiredsystem performance is typically optimized by allocating power at thetransmitter to each subchannel depending on its quality. For example,MIMO power loading typically focuses on optimizing data rate, totaltransmit power, bit error rate, or energy efficiency. However, in oneaspect of the invention, a pattern of subchannel power allocationsprovides for conveying data. For example, a ratio of power allocationsbetween subchannels may be selected to convey information.

FIG. 21 is a block diagram of a receiver configured in accordance withsome aspects of the disclosure wherein the receiver is configured toseparate a clean reference signal from an information-bearing referencesignal and then decode the information-bearing reference. The receivercomprises an RF front-end 2100 coupled to a demultiplexer 2101. A firstoutput of the demultiplexer 2101 comprises a canceller 2102 and anequalizer 2103 coupled to a correlator 2104. A second output of thedemultiplexer 2102 comprises a reference estimator 2122, which is alsocoupled to the correlator 2104. The correlator 2104 is coupled to adecoder 2105. A channel estimator 2124 couples to a synthesizer 2123,which is coupled to the reference estimator 2122.

As described above with respect to FIG. 18, an estimate of the referencesignal (produced by reference estimator 2122) and estimated channelmatrix (produced by the channel estimator 2124) can be used tosynthesize a cancellation signal (in the synthesizer 2123), which isthen subtracted from a received signal (from the first output of thedemultiplexer 2101) in the canceller 2102. The resultinginterference-cancelled signal can be equalized by equalizer 2103, whichcan comprise any of the equalization techniques disclosed herein.Decoding the information-bearing precoding matrix may comprise removingthe reference signal (such as may be performed by the correlator 2104).In one aspect of the disclosure, the data-bearing elements of theprecoding matrix are identified by some predetermined function of theestimated channel matrix. Thus, the channel estimates may be employedfor decoding the received signal.

FIG. 22A is a flow diagram of a communication method according to oneaspect of the invention wherein an N_(T)×N_(R) MIMO system with N_(SS)spatial subchannels per subcarrier frequency is provided withinformation-bearing subchannel power allocations. The number N_(SS) ofspatial subchannels may be selected based on various factors, includingchannel estimates, SNR measurements, the number of transmit and receiveantennas, and the number of undesired receivers. In a Cooperative-MIMOsystem, N_(SS) may depend on the number of available cooperating nodesat either or both the transmit side and the receive side of thecommunication link.

A reference signal is generated 2201 from bits that are coded andmodulated. A set of the coded bits are mapped to a constellation symbolx_(i) for each subchannel iε{1, 2, . . . , N_(SS)}. The received signalyεC^(NR) is a linear transformation of the transmitted signal sεC^(NT)plus an additive noise nεC^(NR), y=Hs+w, where HεC^(NR×NT) is the MIMOchannel matrix, and w has i.i.d. complex Gaussian elements withzero-mean and unit variance.

Channel estimation 2202 is performed by both a transmitting node and areceiving node, and may comprise periodically updating the estimatedMIMO channel matrix H. In a Cooperative-MIMO system, channel estimationmay be performed by one or more of the cooperating nodes. In aspects ofthe invention that employ Cooperative-MIMO, it is understood thatoperations typically performed at each of the transmitting node and thereceiving node may be performed by one or more cooperating nodes. Boththe transmitting node and the receiving node may generate a secret key2203 from their channel estimates.

Both the transmitting node and the receiving node perform SVDfactorization 2204 of their estimated channel H. The SVD factorizationof H is expressed by H=UΣV^(†), where U is an N_(R)×N_(R) unitary matrixwhose columns are the eigenvectors of the matrix HH^(†), V is aN_(T)×N_(T) unitary matrix whose columns are the eigenvectors of thematrix H^(†)H, and Σ an N_(R)×N_(T) diagonal matrix whose diagonalelements are the non-negative real singular values given by[Σ]_(i,i)=√{square root over (λ_(i)(HH^(†)))} for i=1, 2, . . . , r,where λ_(i)(HH^(†)) is the i^(th) largest eigenvalue of matrix HH^(†),and r is the rank of H.

At the transmitting node, transmit power values for each spatialsubchannel are calculated 2205 based on the calculated eigenvaluesλ_(i), the secret key, and an information signal to be conveyed to thereceiving node. The resulting coded power allocation may take the formof a power matrix, P.

The transmitted signal s is generated 2206 via a linear transformations=VPx, where V is an N_(T)×N_(SS) transmit beamforming/precoding matrixobtained from the SVD of H 2204, P is an N_(SS)×N_(SS) diagonal matrixwhere [P]_(i,i)=P_(i) for i=1, 2, . . . , N_(SS), where P_(i) is thetransmit power allocated to the i^(th) spatial subchannel 2205, andxεA^(NSS) is the reference symbol vector drawn from unit-energyconstellation set A.

FIG. 22B is a flow diagram of a communication method according to anaspect of the invention wherein a receiver of an N_(T)×N_(R) MIMO systemwith N_(SS) spatial subchannels per subcarrier frequency is configuredfor receiving and decoding information-bearing subchannel powerallocations. The reference signal generated at the transmitting node isproduced at the receiving node 2211. For example, the reference signalmay be transmitted to the receiving node, and the receiving node mayproduce 2211 an estimate of the transmitted reference signal.Alternatively, the reference signal may originate from a predeterminedsource (e.g., a signal broadcast by an alternative communication systemthat is received and estimated by both the transmitting node and thereceiving node). In one aspect of the invention, the reference signalmay comprise a predetermined data sequence known by both thetransmitting and receiving nodes prior to establishing the communicationlink and stored in memory at the receiving node. In another aspect ofthe invention, the reference signal may be generated locally by thereceiving node, such as from an algorithm that is common to both thetransmitting node and the receiving node.

The receiving node produces channel estimates 2212 that correspond tochannel estimates produced by the transmitting node 2202. The receivingnode may generate a secret key 2213 that preferably is identical to thesecret key generated by the transmitting node 2203. The receiving nodeperforms SVD factorization 2214 of its estimated channel H.

The received signal y is linearly processed 2215 with an N_(SS)×N_(R)matrix, U^(†), obtained from the SVD of H 2214 to yield {tilde over(y)}=U^(†)y=ΣPx+{tilde over (w)}, where Σ=diag(√{square root over (λ₁)},√{square root over (λ₂)}, . . . , √{square root over (λ_(N) _(SS) )}),and {tilde over (w)}=U^(†)w is an equivalent noise vector with i.i.d.complex Gaussian elements with zero-mean and unit variance. Since thereference signal x is known or estimated, it may be removed from thereceived signal.

The effective receiver SNR for subchannel i is γ_(i)P_(i), where γ_(i)is the receiver subchannel-to-noise ratio of subchannel i and is definedas γ_(i)=γ_(i)=λ_(i)/σ_(i) ², where σ_(i) ² is the variance of the noiseand interference experienced by subchannel i. The subchannel powersP_(i) are decoded 2216 using a predetermined algorithm, which is basedon the calculated eigenvalues λ_(i) and the secret key, to produce anestimate of the transmitted information.

It should be understood that various aspects of the invention may beimplemented in hardware, firmware, software, or combinations thereof. Insuch aspects, any of the steps 301-311 can be implemented in hardware,firmware, and/or software to perform the functions of the presentinvention. For example, the same piece of hardware, firmware, or moduleof software may perform one or more of the steps 301-311. Alternatively,one or more of the steps 301-311 may be performed across multipleplatforms, such as in a distributed computing environment.

When implemented in software, firmware, middleware or microcode, theprogram code or code segments to perform the necessary tasks may bestored in a machine-readable medium such as a storage medium, or memory.A processor(s) may perform the necessary tasks. A code segment mayrepresent a procedure, a function, a subprogram, a program, a routine, asubroutine, a module, a software package, a class, or any combination ofinstructions, data structures, or program statements. A code segment maybe coupled to another code segment or a hardware circuit by passingand/or receiving information, data, arguments, parameters, or memorycontents. Information, arguments, parameters, data, etc. may be passed,forwarded, or transmitted via any suitable means including memorysharing, message passing, token passing, network transmission, etc.

As disclosed herein, the term “computer-readable media” may representone or more devices for storing data, including read only memory (ROM),random access memory (RAM), magnetic RAM, core memory, magnetic diskstorage mediums, optical storage mediums, flash memory devices and/orother machine readable mediums for storing information. The term“computer-readable medium” includes, but is not limited to portable orfixed storage devices, optical storage devices, wireless channels andvarious other mediums capable of storing, containing or carryinginstruction(s) and/or data.

The foregoing description of the specific embodiments so fully revealsthe general nature of the invention that others can, by applyingknowledge within the skill of the relevant art(s) (including thecontents of the documents cited and incorporated by reference herein),readily modify and/or adapt for various applications such specificembodiments, without undue experimentation, without departing from thegeneral concept of the present invention. Such adaptations andmodifications are therefore intended to be within the meaning and rangeof equivalents of the disclosed embodiments, based on the teaching andguidance presented herein. It is to be understood that the phraseologyor terminology herein is for the purpose of description and not oflimitation, such that the terminology or phraseology of the presentspecification is to be interpreted by the skilled artisan in light ofthe teachings and guidance presented herein, in combination with theknowledge of one skilled in the relevant arts.

The invention claimed is:
 1. A non-transitory computer readable storagemedium residing in a transmitter and having computer readable codethereon configured to: select a decoy data signal as a carrier signal;synthesize a precoding matrix comprising data-bearing synthesizedmultipath distortions; and precode the decoy data signal with theprecoding matrix to produce a precoded data signal to be transmitted toan intended receiver.
 2. The medium recited in claim 1, furthercomprising instructions to communicate an undistorted version of thedecoy data signal to the intended receiver.
 3. The medium recited inclaim 1, wherein the synthesized multipath distortions are a function ofchannel measurements made by at least one of the transmitter and theintended receiver.
 4. The medium recited in claim 1, wherein the decoydata signal comprises at least one of a retransmitted signal, a datasignal supplied by a third party, a predetermined data sequence that islocally reproduced by the intended receiver, a predetermined datasequence stored in memory at the intended receiver, a known trainingsequence, and a data sequence generated at the transmitter wherein aclean replica is supplied to the intended receiver via an alternativechannel.
 5. The medium recited in claim 1, wherein the precoding matrixis configured to pre-equalize natural distortions in at least onechannel between the transmitter and the intended receiver.
 6. The mediumrecited in claim 1, wherein the decoy data signal comprises an OFDMsignal, and the medium further comprises instructions to perform adiscrete Fourier transform to generate at least one of the decoy datasignal and the precoded data signal.
 7. The medium recited in claim 1,further comprising instructions to multiplex the decoy data signal withthe precoded data signal.
 8. A non-transitory computer readable storagemedium having computer readable code thereon configured to: receive froma multipath channel, a distorted signal comprising a decoy data signalprecoded with a precoding matrix comprising synthesized data-bearingmultipath distortions; receive a clean decoy data signal; and removenatural multipath distortions and the decoy data signal from thedistorted signal for decoding transmitted data from the synthesizeddata-bearing multipath distortions.
 9. The medium recited in claim 8,wherein the clean decoy data signal comprises at least one of a decoydata signal received from a third-party network, a decoy data signalreceived from a different channel in a network from which the distortedsignal is received, a known training sequence, a locally stored datasequence, and a locally generated data sequence.
 10. The medium recitedin claim 8, wherein the precoding matrix is configured to equalize thenatural multipath distortions such that the distorted signal comprisesthe decoy data signal and the synthesized data-bearing multipathdistortions.
 11. The medium recited in claim 8, wherein the decoy datasignal comprises an OFDM signal, and the medium further comprisesinstructions to perform a discrete Fourier transform of the distortedsignal.
 12. A transmitter, comprising: a distortion synthesizerconfigured to generate a precoding matrix from a data signal to betransmitted to an intended receiver; and a precoder configured to employthe precoding matrix to precode a decoy data signal to produce aprecoded signal comprising information-bearing synthesized channeldistortions.
 13. The transmitter recited in claim 12, further comprisingat least one of a decoy data signal generator configured to produce thedecoy data signal, and a receiver configured to receive a third-partytransmission to employ as the decoy data signal.
 14. The transmitterrecited in claim 12, wherein the distortion synthesizer configures theprecoding matrix to pre-equalize natural distortions in at least onechannel between the transmitter and the intended receiver.
 15. Thetransmitter recited in claim 12, further comprising a secondarytransmitter configured to transmit a clean version of the decoy datasignal to the intended receiver.
 16. The transmitter recited in claim12, wherein the decoy data signal comprises an OFDM signal, and theprecoder comprises a discrete Fourier transform configured to generateat least one of the decoy data signal and the precoded data signal. 17.A receiver, comprising: a receiving system configured to receive from amultipath channel, a distorted signal comprising a decoy data signalprecoded with a precoding matrix comprising synthesized data-bearingmultipath distortions; the receiving system further configured toreceive a clean decoy data signal; and a decoding system configured toremove the decoy data signal and any natural multipath distortions fromthe distorted signal to decode data from the synthesized data-bearingmultipath distortions.
 18. The receiver recited in claim 17, wherein thereceiving system is configured for at least one of receiving the cleandecoy data signal from a third-party network, receiving the clean decoydata signal from a different channel in a network from which thedistorted signal is received, receiving the clean decoy data signal froma locally stored data sequence, and receiving the clean decoy datasignal from a locally generated data sequence.
 19. The receiver recitedin claim 17, wherein the decoding system comprises an equalizerconfigured to equalize the natural multipath distortions in thedistorted signal.
 20. The receiver recited in claim 17, wherein at leastone of the receiving system and the decoding system comprises a discreteFourier transform.
 21. The receiver recited in claim 17, the decodingsystem is configured to employ an encryption key based on measurementsof at least one channel between the receiver and an intended transmitterfor decoding the synthesized data-bearing multipath distortions.